Multi-user acquisition procedure for point-to-multipoint synchronous CDMA systems

ABSTRACT

Disclosed are methods for acquiring a forward channel in a point-to-multipoint CDMA system. One method includes the steps of (a) despreading a received CDMA signal with a first pn code that is known to be present in the received CDMA signal and obtaining a first measure of received signal level; (b) despreading the received CDMA signal with a second pn code that is known not to be present in the received CDMA signal and obtaining a second measure of received signal level; and (c) synchronizing to a desired channel using a difference between the first and second signal levels. In this method the first step of despreading despreads a continuously transmitted side-channel, and the step of synchronizing synchronizes to the side-channel. Also, the step of obtaining a first measure of received signal level obtains a correlation peak, and the step of obtaining a second measure of received signal level obtains a correlation null. Preferably, the steps of despreading and obtaining are accomplished iteratively over a range of n first pn code phase states and over a range of n second pn code phase states.

CROSS-REFERENCE TO A RELATED PATENT APPLICATION

This patent application is a continuation of commonly assigned U.S.patent application Ser. No. 08/606,285, filed Feb. 23, 1996, entitled "AMULTI-USER ACQUISITION PROCEDURE FOR MULTIPOINT-TO-POINT SYNCHRONOUSCDMA SYSTEMS", now U.S. Pat. No. 5,805,584, by S. Kingston et al.

FIELD OF THE INVENTION

This invention pertains generally to code division, multiple access(CDMA) communication systems and, in particular, to direct-sequence (DS)point-to-multipoint synchronous CDMA communications systems.

BACKGROUND OF THE INVENTION

In a CDMA communications system a plurality of user communicationsignals can be transmitted within, i.e., share, a same portion of thefrequency spectrum. This is accomplished by providing a plurality ofdifferent pseudonoise (pn) binary code sequences (e.g., one for eachuser) that modulate a carrier, thereby "spreading" the spectrum of theresulting waveform. In a given receiver all of the user signals arereceived, and one is selected by applying an assigned one of the pnbinary code sequences to a correlator to extract only the signal energyintended for the receiver, thereby "despreading" the received CDMAtransmission. All other (uncorrelated) user transmissions appear asnoise.

One type of CDMA communication system is specified by a documentreferred to as EIA/TIA/IS-95. The system as specified uses a pluralityof base stations that establish and maintain bidirectionaldirect-sequence (DS) CDMA links with a plurality of mobile stations(e.g., cellular telephones). One feature of the IS-95 system is thepresence of a pilot channel that is transmitted by each base station.

The pilot channel is an unmodulated, direct-sequence spread spectrumsignal that is transmitted continuously by each CDMA base station. Thepilot channel allows a mobile station to acquire the timing of theForward CDMA channel (i.e., from the base station to the mobilestation), provides a phase reference for coherent demodulation, andprovides a reference for signal strength comparisons between basestations for determining when to handoff. The pilot pn sequence isdefined as a pair of modified maximal length PN sequences with period2¹⁵ that are used to spread the Forward CDMA channel and the ReverseCDMA channel. Different base stations are identified by different pilotPN sequence offsets. A pilot pn sequence offset index is defined to bein units of 64 pn chips, relative to a zero offset pilot pn sequence. Apn chip is defined as one bit in the pn sequence. The pilot strength isdefined as the ratio of received pilot energy to overall receivedenergy.

Walsh functions are a class of 2^(N) time orthogonal binary functionsthat are used to establish orthogonality between the different pn binarycode sequences used by the pilot and user channels.

The use of the pilot channels, while providing certain advantages in aCDMA system intended for use with mobile stations, may presentdisadvantages as well, particularly in systems where the usertransceivers are fixed as opposed to mobile. For example, the pilotchannels consume some amount of the available pn code sequences andsignal energy, all of which could be otherwise allocated to the users ofthe system.

Also, in many detection approaches synchronization to the pn code timingmust be achieved before carrier phase-lock can occur. In this case anon-coherent detection algorithm must be employed. Generally,non-coherent detectors rely on energy detection within a fixedbandwidth, as a range of code timing cells are searched. Upon locatingthe correct code timing, the detector energy level rises above apredetermined threshold level. A bit-sync loop then takes over to obtainthe finer-resolution bit timing.

However, standard acquisition approaches are known to fail when thenumber of users becomes large. This is due to the fact that the noisepower becomes comparable to the signal power when the user of asynchronous type of CDMA system is not synchronized. As a result, itbecomes very difficult for the user's receiver to distinguish thecorrect pn timing phase from the incorrect phases resulting from theincreased noise.

As can be appreciated, the acquisition technique is an important aspectof the receiver, in that its operation impacts the overall speed atwhich synchronization to the forward link occurs. If the time requiredto synchronize the user's receiver becomes excessive, the delay may beconsidered as objectional by the user.

OBJECTS OF THE INVENTION

It is a first object of this invention to provide methods and apparatusto enable a receiver to synchronize to a synchronous CDMA communicationssystem.

It is a further object of this invention to provide methods andapparatus for implementing a synchronous CDMA system wherein a firstforward channel transmits a null (inactive) pn code sequence that isorthogonal to all active codes, a second forward channel transmits analways-active pn code sequence, and wherein a receiver uses either thefirst forward channel, the second forward channel, or both of theforward channels to synchronize to the forward CDMA link.

SUMMARY OF THE INVENTION

The foregoing and other problems are overcome and the objects of theinvention are realized by methods and apparatus in accordance withembodiments of this invention, wherein a user terminal is provided withcircuitry and methods enabling the acquisition of a desired CDMA channelin the presence of a plurality of synchronous interfering channels.

In one aspect, this invention teaches a method for synchronizing to aforward channel in a CDMA system. The method includes steps of (a)despreading a received CDMA signal with a first pn code that is knownnot to be present in the received CDMA signal, (b) obtaining a measureof received signal level, and (c) using the obtained measure of thereceived signal level when setting a phase of a second pn code thatcorresponds to a desired forward channel that is to be received. In thismethod the step of despreading despreads a null channel, and the desiredforward channel is a continuously transmitted side-channel that providessystem-level information to all subscriber units or terminals.

In another aspect this invention teaches a method for acquiring aforward channel in a point-to-multipoint CDMA system. This methodincludes the steps of (a) despreading a received CDMA signal with afirst pn code that is known to be present in the received CDMA signaland obtaining a first measure of received signal level; (b) despreadingthe received CDMA signal with a second pn code that is known not to bepresent in the received CDMA signal and obtaining a second measure ofreceived signal level; and (c) synchronizing to a desired channel usinga difference between the first and second signal levels.

In this method the first step of despreading despreads a continuouslytransmitted side-channel, and the step of synchronizing synchronizes tothe side-channel. Also, the step of obtaining a first measure ofreceived signal level obtains a correlation peak, and the step ofobtaining a second measure of received signal level obtains acorrelation null.

Preferably, the steps of despreading and obtaining are accomplishediteratively over a range of n first pn code phase states and over arange of n second pn code phase states. For a pn code phase state i ofthe plurality n of first and second pn code states, the method furtherdetermines a difference value between the first measure of receivedsignal level and the second measure of received signal level; comparesthe difference value to a threshold value; and, if the difference valueis greater than the threshold value, sets a pn code generator to outputa pn code corresponding to the desired channel, wherein the outputted pncode is set at the pn phase state i. Else, if the difference value isnot greater than the threshold value, the method instead increments thepn phase state i and re-executes the steps of despreading and obtaining.

In this method the step of determining the difference value includes astep of storing the determined difference value, and if none of the ndifference values are greater than the threshold value, the methodfurther includes a step of examining the stored difference values toselect a stored difference value that has a largest value; and a step ofsetting the pn code generator to output the pn code corresponding to thedesired channel, wherein the outputted pn code is set to a pn phasestate that corresponds to the selected stored difference value.

In a further aspect, this invention teaches a synchronous CDMAcommunication system that operates in accordance with the foregoingmethods.

BRIEF DESCRIPTION OF THE DRAWINGS

The above set forth and other features of the invention are made moreapparent in the ensuing Detailed Description of the Invention when readin conjunction with the attached Drawings, wherein:

FIG. 1 is a simplified block diagram of a synchronous, DS-CDMAcommunications system that is constructed and operated in accordancewith this invention, the system having a radio base unit (RBU) and aplurality of subscriber units (SUs).

FIG. 2 is a block diagram of a first embodiment of the SU receiverdetector of FIG. 1, in particular a non-coherent square law detector.

FIG. 3 is a block diagram of a second embodiment of the SU receiverdetector of FIG. 1, in particular a non-coherent absolute valuedetector.

FIG. 4 is a graph that illustrates the relative mean acquisition timeperformance of the single-user, multi-user, and difference tests of thisinvention, for the case E_(s) /N_(o) =6 dB, P_(d) =0.995, and α-0.01,and for a range of numbers of active users.

FIG. 5 is a graph that illustrates an average multi-user interferencepower and average side-channel correlation power versus code timingoffset, with 3 pole Butterworth transmitter and receiver filters, and 30active users.

FIG. 6 is a graph that illustrates energy versus offset for a desiredsubscriber unit, for multi-user interference, and for background noise.

FIGS. 7A-7D are graphs that illustrate energy versus offset curves forthe outputs of matched filters that are matched to the side-channel PNcode (dashed line) and a null-code (solid line) for the cases of highand low SNR, as well as heavy and light loading. In these Figures thesymbol X=matched filter output for the side-channel code, and the symbol□=matched filter output for the null-code.

FIG. 8 is a logic flow diagram of a channel acquisition method inaccordance with this invention.

DETAILED DESCRIPTION OF THE INVENTION

Referring to FIG. 1, a synchronous CDMA communications system 10, whichin presently preferred embodiments of this invention is embodied as afixed wireless system (FWL), is considered herein to be a CDMA systemwherein forward link (FL) transmissions from a radio base unit (RBU) 12for a plurality of user or subscriber units (SUs) 14 are bit and chipaligned in time, and wherein the SUs 14 operate in accordance with theteaching of this invention for receiving the FL transmissions and forsynchronizing to one of the transmissions. The FWL is suitable for usein implementing a telecommunications system that conveys voice and/ordata between the RBU 12 and the SUs 14.

The RBU 12 includes circuitry for generating a plurality of user signals(USER₋₋ 1 to USER₋₋ n), a side channel (SIDE₋₋ CHAN) signal that iscontinuously transmitted, and a NULL signal. Each of these signals isassigned a respective pn spreading code and is modulated therewithbefore being applied to a transmitter 12a having an antenna 12b. Whentransmitted on the FL the transmissions are modulated in phasequadrature, and the SUs 14 are assumed to include suitable phasedemodulators for deriving in-phase (I) and quadrature (Q) componentstherefrom. The illustrated arrangement is for one frequency (carrier)channel, it being realized that the RBU 12 is capable of transmitting aplurality of such frequency channels. By example, each frequency channelincludes up to 31 code channels, and has a center frequency in the rangeof 2 GHz to 3 GHz.

Each SU 14 includes an antenna 14a, a mixer 14b for down-converting thereceived signal, a correlator 14c wherein the user's transmission isobtained by despreading the received signal with a local pn code, and adetector and correlator 14d. Suitable embodiments for the detector are anon-coherent square law detector shown in FIG. 2 and also a non-coherentabsolute value detector shown in FIG. 3. The SU 14 also includes a localprocessor 14e that is responsible for managing the operation of the SU14. These management functions include generating a variable localoscillator (LO) signal, such as is obtained from a voltage controlledoscillator (VCO) 14f, and providing the pn binary code sequence that isassigned to the SU 14 for despreading the user's signal. The processor14e is also responsible for executing one or more of the acquisitionmethods in accordance with this invention. Although the SU 14 is capableof also transmitting a DS-CDMA signal to the RBU 12 on a return link,these functions are not germane to the teaching of this invention andare thus not illustrated.

For the presently preferred embodiments of this invention the antennas12b and 14a have a line-of-sight relationship, the SUs 14 are fixed inlocation with respect to the RBU 12, and the antennas 12b and 14a areboresighted during installation of the SU 14. However, and as will bediscussed below, the teachings of this invention are not limited to onlythis particular presently preferred arrangement.

The ensuing description assumes the use of a DS signal, a(t), with codesymbol duration T_(s), multiplied by a spreading sequence c(t), withchip duration T_(c) and a null-to-null bandwidth W_(c) =2/T_(c). Thesymbol rate for each SU 14 is fixed at 1/T_(s), and the chipping rate at1/T_(c) =P/T_(s). All of the pn codes are mutually orthogonal whenaligned, and are assumed to be accurately aligned during normaloperation. In the presently preferred embodiments of this invention thepn codes are selected from a set of randomized Walsh-Hadamard codes. Theteaching of this invention is not, however, limited to only signalshaving these characteristics. By example, the set of pn spreading codescan be selected from any set that exhibits low cross-correlation at zerorelative shift.

In the absence of multi-user interference, the received signal can bewritten as r(t)=a(t-r)c(t-r)+n(t), where n(t) represents white gaussianchannel noise. r(t) is multiplied by successive shifts of the spreadingsequence, c(t-mδt), in an attempt to estimate the timing offset, r. Thedetector filter 17 of FIGS. 2 and 3 is a finite (T_(s)) integrator withan equivalent noise bandwidth W_(s) =1/T_(s). Within this bandwidth, theeffective noise power spectral density, 1/2N_(o), is unchanged by theintegrator. When c(t-mδt) is synchronous with r(t); the pn spreadingcode is collapsed, and the integrator produces mean values of the DSsignal, a(t-r). Assuming non-synchronous operation, most of the signalenergy falls outside the bandwidth of the detector filter 17 and, to agood approximation, the detector input can be taken as white gaussiannoise only. For worst case non-synchronous operation, the signal energyis reflected back into the spread bandwidth, W_(c).

The sum of n samples of a squared gaussian process is central χ²distributed with n degrees of freedom, and has a bandwidth (near DC)twice that of the detector filter 17. The central χ² distribution with ndegrees of freedom can be written ##EQU1## where σ is the variance ofthe gaussian process.

Given spread code synchronization (i.e., mδt≈r), the spectrum of thereceived signal is collapsed to the data bandwidth, thus increasing thetotal energy within the detector filter 17. Because the phase of thereceived signal rotates freely with respect to the reference, thiswaveform is not DC, even for a constant energy signal. It would thusappear that the output of the detector should again be central χ²distributed, but with a larger mean value than for the noise only case.However, the phase rotation rate is assumed to be slow enough that thesignal component can be treated as deterministic, and hence the ithindependent sample of the detector filter output is distributed normallywith mean s_(i) and variance σ² determined by the noise component. Theintegrator output is then characterized by a non-central χ² distributionwith n degrees of freedom, and a non-centrality parameter: ##EQU2##where S represents the average signal power. The non-central χ²distribution with n degrees of freedom can be written as: ##EQU3##

Finally, a summation of n samples from block 21 is compared in block 23to a preset threshold, t_(h). The threshold and the number of samplesare set to achieve a probability of detection ≧P_(d) simultaneously witha probability of false alarm ≦α. Give t_(h), P_(d) can be determined byintegrating the non-central χ² distribution from t_(h) to ∞. Theprobability of false alarm, α, is determined by integrating the centralχ² distribution from t_(h) to ∞. As the integration time, T, becomeslarger, the spread between the two distributions also becomes larger. Atsome point, the required detection probability can be achieved whilesimultaneously achieving the target false alarm probability. The minimumnumber of samples required to achieve the target detection and falsealarm probability criteria determines the observation window of thedetector, T=nT_(s).

However, normally some time uncertainty will exist, and some number (Q)of time cells must be searched. The mean acquisition time is thendetermined as follows: Given a H_(o) case (a noise only case), the timerequired to search a single time cell is given by the expression

    T.sub.o =T+αKT,

where K is referred to as the cost for a false alarm. If it is assumedthat, upon exceeding the threshold, a verification procedure is begun,and that this verification procedure includes a second integration oflength KT, which is long enough to assure that the probability of asecond false alarm is negligible, then T_(o) =(1+αK)T. On average,(1/2Q-1) time cells must be searched before encountering the correctcell. Next, the time required to search a time cell given by a H₁ case(a signal present case), is given by the expression

    T.sub.1 =P.sub.d (1+K)T+(1-P.sub.d) (1/2QT.sub.o +T.sub.acq).

The first term corresponds to a successful detection, in which case averification procedure is also required. The second term corresponds toa detection failure (in which case the acquisition procedure must bestarted from the beginning, but increased by 1/2QT_(o) (since all timecells must now be searched). The acquisition time is therefore given bythe expression

    T.sub.acq =(1/2Q-1)T.sub.o +T.sub.1.

Substituting and rearranging, one arrives at the expression ##EQU4##

Hence, given the signal-to-noise ratio (SNR), the required detectionprobability, P_(d), the false alarm probability, α, and a cost factor,one can compute the necessary integration time, T. Then, given T and theuncertainty, Q, the mean acquisition time, T_(acq), can be determined.

For a particular value of E_(s) /N_(o) at the input filter, the inputsignal-to-noise ratio is given by ##EQU5##

Assuming that the signal is AGC controlled at the output of the SUreceiver filter 15, then S+N_(c) =1, from which it follows that ##EQU6##where S is the signal power, and N_(c) the noise power at the output ofW_(c). In addition, N_(o) =2N_(c) /W_(c). The noise power at the outputof the detector filter 17, W_(s), is then N_(s) =N_(o) W_(s) /2, fromwhich the signal-to-noise ratio in the energy detector 19a or 19b can becalculated as SNR_(s) =S/N_(s). The probability distribution functionsfor each hypothesis are then completely determined by the parameters:

    σ.sub.0.sup.2 =σ.sub.1.sup.2 =N.sub.s and λ=nS.

The foregoing analysis is now generalized to a multi-user channel ofmost interest to this invention. By example, there may be M≦30 activeusers (SUs 14) in the CDMA communication system 10, each of whichreceives coded information symbols from the RBU 12, with an assignedlength P=32 code. All of the pn codes are mutually orthogonal whenaligned, and are assumed to be accurately aligned during normaloperation. The symbol rate for each SU 14 is fixed at 1/T_(s), and thechipping rate at 1/T_(c) =P/T_(s). The RBU 12 transmits all activechannels together synchronously and uniformly, and hence the channelpower levels and timing offsets received at any one SU 14 aresubstantially equal.

In addition to the user channels, there are two additional channelswhich all SUs 14 make use of. One, which is never active, is referred toas the above-described null-channel. Although the null-channel isassigned a unique pn code (referred to herein as the null-code), whichis orthogonal to every active code, the null-code is not actuallytransmitted. That is, the null-code can be considered as a "missing"code. This is schematically shown in FIG. 1 by the open switch (SW)placed in the NULL signal path. The switch (SW) could also be placed inthe pn₋₋ null path, such that the pn code assigned to the null channel(pn₋₋ null) does not reach the associated multiplier (spreader) 12c.

The second channel, referred to above as the side-channel, is alwaysactive. The side channel provides side-information to the SUs 14 (e.g.,system access information, etc.). The side-channel is used as well forcode-synchronization, as described below.

In a single-user acquisition technique or test (SUT), only aside-channel statistic is employed, while in a multi-user acquisitiontechnique or test (MUT), the null-channel statistic is employed. It isalso within the scope of this invention to employ two differentstatistics; one corresponding to the null-channel and the other to theside-channel, as explained below.

Single-user technique: For this measurement, the side-channel pn code isused as the SU 14 correlation code, c(t-m∂t), as shown in thenon-coherent square law detector block diagram of FIG. 2 and also thenon-coherent absolute value detector block diagram of FIG. 3. Thesingle-user technique makes use of this statistic alone in an attempt tolocate a correlation peak between the copy of the side-channel pn codestored in the SU 14 and the active side-channel itself. The maximumpoint on this peak occurs when the codes are aligned.

The multi-user technique: For this measurement, the null-code is used asthe correlation code. The multi-user test makes use of this statisticalone in an attempt to find a correlation null (as opposed to peak)between the copy of the null-code stored in the SU 14 and the activechannels. The correlation null results from the mutual orthogonality ofevery possible active code with the null-code, and coincides with codealignment.

It has been found that the single-user test may be reduced ineffectiveness when M approaches the maximum number of allowed activeusers. This is due to the fact that the multi-user interference leveldrops just as the side-channel correlation signal rises, in a nearlyself-canceling fashion. Similarly, when M is small, it has been foundthat the multi-user test may be reduced in effectiveness, because thecorrelation null can be masked by ambient noise.

In order to compensate for these effects, it is preferred to use adifference between these two statistics when acquiring the FL. Hence,for large M, the single-user statistic dominates, and for small M, themulti-user statistic dominates. But, in either case, the differencebetween the two statistics changes significantly (i.e., becomes larger)as the pn code timing approaches the lock point. This advantage ismitigated somewhat by the doubling of an observation window, T=2nT_(s).However, and referring now to FIG. 4, it can be readily seen that theuse of the difference test provides an overall advantage in meanacquisition time when compared over the full range of possible numbersof active users.

The detection in accordance with this invention is thus preferablyperformed on two probability distribution functions (pdf) resulting fromthe difference between the two statistics described above. As in thesingle-user detection case, one pdf corresponds to a H_(o) case, orcodes mis-aligned case, the other to a H₁ case, or codes aligned case.In either case, and because the two statistics are computedsequentially, the noise components are essentially uncorrelated. The pncodes are length P=32, and the timing is searched in the SU 14 inhalf-chip increments (δt=T_(c) /2), thus giving Q=64 possible positions.The time cell yielding the best signal level will therefore, at worst,be off by 1/4T_(c).

Given the H_(o) case, both the single-user and multi-user tests yieldambient gaussian white noise and multi-user interference. Since themulti-user codes are uncorrelated with the test code, both noisecomponents are assumed gaussian, and hence central a χ² distributed atthe detector output. The channel noise variance is N_(s) =1/2N_(o)W_(s), where 1/2N_(o) =N_(c) /W_(c) is the two-sided noise spectralpower density. Hence N_(s) =N_(c) T_(c) /2T_(s) =1/2N_(c) /P. The worstcase multi-user interference power is I_(s) =1/2I_(o) W_(s), where1/2I_(o) is the two-sided interference spectral density.

It should be noted that the interference density varies over thenull-to-null bandwidth of the receiver filter 15 of FIGS. 2 and 3. Ofmost interest, however, is the density near DC. The equivalentinterference bandwidth at this power density is 1/2W_(c) =1/T_(c).Hence, 1/2I_(o) =I_(c) T_(c), and I_(s) =I_(c) T_(c) /T_(s) =I_(c) /P.For the H_(o) hypothesis, then σ₀ ² =I_(s) +N_(s) +S/P=1/p 1-1/2N_(c) !,where S/P represents the contribution of the side-channel.

Given the H₁ case, the single-user test yields the ambient whitegaussian noise term and a deterministic signal component, and hencecontributes a non-central χ² distributed random variable at the outputof the detector 12d. The multi-user test yields ambient white gaussiannoise which contributes a central χ² distributed random variable. Thisassumes, of course, that the multi-user interference can be ignored dueto the mutual orthogonality of the aligned pn spreading codes. However,and because of filtering, the pn codes are typically not strictlyorthogonal. It can be shown that, with 3-pole butterworth filtering, theinterference null depth is approximately 25 dB upon pn code alignment.Thus, for worst case fades (E_(s) /N_(o) ≈3 dB), the interference nullcan be well below the ambient white gaussian noise level. Another factorto consider results from the fact that the worst case alignment offsetof 1/4T_(c) will have the effect of weakening the interference null andthe signal correlation peak. From the graphs shown in FIG. 5, themulti-user correlation null can be seen to lose as much 14 dB and thesingle-user correlation peak about 2 dB. Thus, these effects are takeninto consideration. It can be assumed however, that for a worst casefade, the noise variance is approximately σ₁ ² =N_(s) for both thesingle-user and multi-user statistics. The non-centrality parameter λ isequal to nS.

The probability distribution function for a random variable which is thedifference between two other independent random variables, Z=X-Y, can bedetermined from the integral: ##EQU7## which is similar to a convolutionintegral. The H_(o) and H₁ distributions can thus be obtained by directintegration, or from the inverse fourier transform of the product ofmodified forms of the characteristic functions for the central andnon-central χ² distributions.

For the multi-user acquisition case, the SU 14 automatic gain control(AGC) function is assumed set such that S+N_(c) +I_(c) =1, where I_(c)=MS is the multi-user interference power. Hence, (M+1)S+N_(c) =1. As isalso the case for the single-user approach, the SNR can be defined as:##EQU8##

It thus follows that: ##EQU9##

Upon setting λ=nS, the probability distributions are completelydetermined up to the number of degrees of freedom, which is adjusted tomeet the probability of detection and false alarm criteria: ##EQU10##for some threshold t_(h).

Based on the foregoing, and in accordance with a first aspect of thisinvention, there is provided a method for the SU 14 to acquire thecorrect code timing for a synchronous CDMA forward channel link in thepresence of some amount of multi-user interference (MUI). This methodexploits the orthogonality of the pn codes used to determine the propercode phase by purposely despreading a pn code (i.e., the null code)which is known to not be present. This is done to overcome the problemthat arises when using a conventional acquisition procedure when manyusers are active. In this case the interference energy of P-1interfering users, with processing gain P, can be nearly as strong asthe energy of the desired user's signal. As a result, the standardacquisition approach, wherein a detection is declared when the energy ofa matched filter output (or sliding correlator output) is larger than athreshold, is not viable in a heavily loaded system.

To simplify the discussion of the method, it is assumed that thefrequency offset between the carrier of the incoming signal and thereceiver's local oscillator is zero.

FIG. 6 illustrates the energy of the matched filter output due to thedesired user's signal, due to the interfering users, and due to thebackground noise as a function of the timing offset of the matchedfilter. It should be noted that on the forward channel, the user signalsare all assumed to perfectly synchronized with each other. The differentcurves shown in FIG. 1 represent the various components of the receivedsignal, and the sum of these components make up the received strength.It is important to note that the MUI energy due to a receiver chiptiming offset is comparable to the energy of the desired user's signalwhen the receiver is not offset. The implication of this is that astandard acquisition algorithm would not be able to easily distinguishthe difference between the offset and synchronized phases. In fact,unless additional averaging takes place, the signal energy isessentially equal to the noise variance, which implies a detectionsignal-to-noise ratio (SNR) of approximately 0 dB.

It should also be noted that the noise due to MUI and the noise due tobackground noise are both zero mean noise processes. In the conventionaldetection procedure, the absolute value of the larger of I and Qdespreader outputs are taken and the results are averaged over somedwell time. The fact that the noise has a zero mean implies that it ispossible to increase the decision SNR by averaging the detectionstatistics. However, in the case of a heavily loaded system, theaveraging time would be required to be long in order to create a largeenough SNR to determine reliably whether the receiver is correctlysynchronized to a signal.

In contrast with the standard acquisition approach, and in accordancewith an aspect of this invention, if the SU receiver instead despreads aPN code which is not transmitted, then the noise will be "tuned out"when the receiver comes into alignment with the interfering signals.This implies that the SU receiver can look for the "hole in the noise"which occurs at the zero offset phase, and thus determine when the noiseenergy at a sliding correlator output drops below a threshold. At thistime the acquisition circuit can declare that a lock has occurred. Thistechnique has been referred to above as the multi-user test (MUT).

Although this approach provides significant advantages over theconventional acquisition technique, two problems can arise during theuse of the MUT approach.

The first problem results from the fact that the number of interferingusers is not constant, and so the absolute level of the MUI is notfixed. This complicates the acquisition problem because in the lightlyloaded case, where the background noise is non-negligible, the noisepower may not decrease at the output of the matched filter when the codealignment becomes perfect.

The second problem results from the fact that the threshold of noisepower at which a lock should be declared is a function of the number ofcurrently active interfering users, and so some type of calibrationprocedure should be performed to set this threshold. Such a calibrationprocedure may involve scanning the possible code phases in half-chipsteps to determine an average energy level and the lowest energy level.If the lowest energy level is found to be lower than the average levelby more than some threshold amount, then the lowest energy level phasemay be declared to be the lock phase. The penalty associated with thisapproach, however, is that the calibration procedure can require asubstantial amount of time to perform, namely 2P times the dwell time.While a lengthy acquisition time for the SU is not a serious drawback,since it only occurs at startup and after long deep fades, it would bepreferable to avoid a relatively long mean time to acquisition ifpossible.

FIGS. 7A-7D illustrate four possible situations of loading (number (N)of active interfering users) and background noise levels. In each case,the matched filter or sliding correlator output is shown, when matchedto the side-channel pn code and to the null pn code (i.e., the pn codethat is not transmitted). It is clear that the behavior of the null pncode and side-channel pn code sliding correlators varies by a largeamount depending on the channel loading and background noise level.

In a presently preferred embodiment of this invention both the signaluser test (SUT) and the multi-user test (MUT) are performed, and theresults are subtracted at each phase state. The procedure for thispresently preferred technique is shown in the logic flow diagram of FIG.8.

In this procedure, at each of the 2P phase states (for example 64 phasestates), and beginning with a phase x (Block A), the SU receiver firstperforms the signal user test (SUT) by averaging the absolute value ofsome number of the larger of the I and Q despreader outputs (Block B).The SU receiver then performs the multi-user test (MUT) at Block C bydoing the same averaging with the despreader employing the null pn code.The difference of these two test statistics is then computed (Block D)and if the difference, Z(x), is larger than a threshold, a detection isdeclared (Block E). If the threshold is not exceeded, the SU processor12e stores Z(x) in an associated memory (MEM). At Block F adetermination is made by the processor 12e if all 64 possible phasestates of phase x have been checked. If no, control passes to Block Gwhere the processor 12e slips the receiver's chip-clock to the nextphase state, and the loop is re-entered at Block B.

If all 64 possible phase states are exhausted and no detection isdeclared at Block F, then there at least two options. The first optionis that the SU processor 12e can sort the stored Z(x) values and pickthe phase state, x, with the largest Z(x) value (Block H). The secondoption is that the receiver can continue slipping the x value throughthe 64 possible states again in the event that a temporary fade wasresponsible for the missed detection on the first pass.

This method of using the relative differences of the two tests (SUT andMUT), rather than an absolute test result, provides a test which is morerobust, as well as a test which yields a lower mean time to acquisitionthan the use of an absolute test.

The overall purpose of the method is to provide a multi-user acquisitionprocedure for use by the SU 14 in acquiring the forward side-channel inthe presence of many interfering forward channel signals. An importantaspect of this method is that by comparing the result of the single usertest (SUT) to that of the multi-user test (MUT), for each possible phasestate, the relative difference between the two tests is better able todistinguish the correct phase state from all of the incorrect phasestates.

The presently preferred embodiment of this invention assumes that notiming (reference) information is present, and up to, by example, 10symbol rates of frequency uncertainty exists. It is also assumed thatall interfering channels transmit independent random data.

Further in accordance with an aspect of this invention, a singleacquisition sequence, also referred to as a "look", includes setting aparticular chip code phase and carrier frequency and thereafteraccumulating a detected value (i.e., a magnitude of an in-phase (I)component plus a magnitude of a quadrature (Q) component) of, nominally,64 symbols. This operation is accomplished by the detector circuitry12d, and the resulting value is supplied to the processor 12e whichcontrols the overall acquisition sequence. By example, at 17 Kbaud thisoperation can occur every 3.8 milliseconds.

The acquisition technique in accordance with this embodiment of theinvention is comprised of four components; referred to herein as channelevaluation, code phase search, carrier frequency pull-in, and carrierlock.

The channel evaluation component operates to search for a region oforthogonality during periods of heavy use (high communications traffic).A pn spreading code for an unused channel (the null channel) is suppliedand a timing search is performed at a nominal carrier frequency in halfchip steps. A reduction in a detected signal energy level is assumed tobe indicative of the presence of a region of orthogonality, i.e., a pncode that is approximately equal to the proper pn timing. If look valuesare found below a threshold proportional to a mean look value, timingdetection is declared. Longer looks at plus and minus 1/2 chip steps andfrequency steps of 1/4 the symbol rate may then be made to more closelydetermine a lock point. The desired channel's (e.g., the side-channel)pn code is then switched in, and the carrier frequency pull-in isimplemented.

If no pn code phase is found to result in the detection of a signalbelow threshold, the carrier frequency is incremented in, by example,half symbol rate steps and the channel evaluation procedure is repeated.The use of half symbol rate steps assumes that orthogonality issignificantly reduced for frequency offsets of 1/4 symbol rate. If thefrequency uncertainty is exhausted without locating a region oforthagonality, it is assumed that insufficient interference sources, orinterferers, exist to produce such a region and a conventional codephase search is performed.

The above-described channel evaluation search uses, by example, 256 pncode phases with, by example, 20 frequency increments and takesapproximately 20 seconds to complete. The channel evaluation searchrequires the use of the channel known to be vacant (i.e., the nullchannel).

The conventional code phase search is performed if the channelevaluation search does not produce a result. The conventional code phasesearch uses the pn code of the channel to be acquired (e.g., theside-channel). The pn code space is searched in half chip steps for eachfrequency in 1/2 symbol rate steps, and seeks to locate a look valueabove a threshold that is proportional to the mean value. A full chipsearch is performed and a largest look value above threshold is taken asthe lock point. As a result, any multipath signal weaker than the directsignal will also be detected. If the conventional code phase search isunsuccessful the channel evaluation search may be tried again.

Once the timing phase is established to within 1/4 chip by one of thetwo above-described code phase search techniques, and the carrierfrequency is tuned to within 1/4 symbol rate, the chip timing loop isclosed and a carrier frequency pull-in is accomplished. The frequencypull-in procedure operates to tune the LO frequency (see FIG. 1) to apoint where phase lock can be rapidly accomplished without false lockingto symbol rate sub-harmonics.

An intra-symbol discriminator may also be employed. The intra-symboldiscriminator uses another pn code channel which has a pn code thatrepresents the desired pn code but with the second half inverted. Achannel using that particular code will interfere with thediscrimination process, but the interference will a zero meaninterference (assuming random data) that is readily filtered by the loopfilter. The frequency pull-in procedure terminates after a fixed time atwhich the SU 14 is switched to phase lock, thereby ending theacquisition procedure. Thereafter the SU 14 continuously performscarrier phase error detection to determine a loss of lock. If a loss oflock is detected the SU 14 returns to the channel evaluation search.

While the invention has thus been particularly shown and described abovewith respect to a number of embodiments thereof, it will be understoodby those skilled in the art that changes in form and details may be madetherein without departing from the scope and spirit of the invention.

For example, it should be realized that the teaching of this inventionis not limited to any of the exemplary frequencies, pn code lengths,numbers of users, despreader and detector embodiments, and the like thatwere described above. Furthermore, it should be realized that thethreshold in Block D of FIG. 8 can be made variable or adaptive as, byexample, a function of the number of times that the method loops throughthis block. An initial setting of the threshold can be accomplished inaccordance with calibration procedure that slips through all phasestates, and sets the threshold to lie between the highest and the secondhighest measured correlation values.

It should further be understood that the teaching of this invention isnot limited for use with the RF transmitters and receivers illustratedin FIG. 1. That is, in other embodiments of this invention the CDMAforward and reverse link signals can be conveyed through, by example,coaxial cable or fiber optic cable. The CDMA signal could also beconveyed through water, using suitable acoustic transducers.

The above described embodiments should thus be viewed as being exemplaryof the teaching of this invention, and should not be construed in alimiting sense upon the practice of this invention.

What is claimed is:
 1. A method for operating a synchronous CDMAcommunications system, comprising steps of:providing a plurality ofsubscriber units (SUs); providing at least one radio base unit (RBU) fortransmitting a signal comprised of a plurality of first forward channelsindividual ones of which being intended for receipt by one subscriberunit (SU), and for transmitting at least one second forward channel thatis intended for receipt by a plurality of SUs, each of said forwardchannels being spread with an associated spreading code that has,ideally, an orthogonal relationship to spreading codes of the otherforward channels; receiving the transmitted signal at a SU; despreadingthe received signal using a spreading code associated with the secondforward channel to obtain a first measure of received signal energy;despreading the received signal using a null spreading code that isknown not to be transmitted by the RBU to obtain a second measure ofreceived signal energy, the null spreading code having, ideally, anorthogonal relationship to the spreading codes of the forward channels;and in the SU, adjusting a phase of the spreading code associated withthe second forward channel based on the first and second measures ofreceived signal energy.
 2. A method as in claim 1, wherein the step ofobtaining a first measure of received signal energy obtains acorrelation peak, and wherein the step of obtaining a second measure ofreceived signal energy obtains a correlation null.
 3. A method as inclaim 1, wherein the steps of despreading and obtaining are performedover a range of phase states.
 4. A method as in claim 3, and furthercomprising steps of:subtracting the results obtained for each phasestate; comparing a difference value to a threshold value; andterminating the steps of despreading and obtaining when the differencevalue exceeds the threshold value, and storing the corresponding phasestate; else if the difference value does not exceed the threshold valueafter performing the steps of despreading and obtaining over the rangeof phase states, storing the phase state corresponding to a largestdifference value; wherein the stored phase state is used in the step ofadjusting.
 5. A synchronous CDMA communications system, comprising:aplurality of subscriber units (SUs); at least one radio base unit (RBU)for transmitting a signal comprised of a plurality of first forwardchannels individual ones of which being intended for receipt by onesubscriber unit (SU), and for transmitting at least one second forwardchannel that is intended for receipt by a plurality of SUs, each of saidforward channels being spread with an associated spreading code thathas, ideally, an orthogonal relationship to spreading codes of the otherforward channels; a receiver in each said SU for receiving thetransmitted signal; a despreader in each said SU for despreading thereceived signal using a spreading code associated with the secondforward channel to obtain a first measure of received signal energy;said despreader further despreading the received signal using a nullspreading code that is known not to be transmitted by the RBU to obtaina second measure of received signal energy, the null spreading codehaving, ideally, an orthogonal relationship to the spreading codes ofthe forward channels; and control means in each said SU for adjusting aphase of the spreading code associated with the second forward channelbased on the first and second measures of received signal energy.
 6. Asynchronous CDMA communications system as in claim 5, wherein thedespreader, when obtaining the first measure of received signal energy,obtains a correlation peak, and when obtaining the second measure ofreceived signal energy obtains a correlation null.
 7. A synchronous CDMAcommunications system as in claim 5, wherein said despreader in eachsaid SU operates over a range of phase states.
 8. A synchronous CDMAcommunications system as in claim 7, wherein said control means in eachsaid SU is operable for subtracting the results obtained for each phasestate and for comparing a difference value to a threshold value; saidcontrol means being responsive to a first condition wherein thedifference value exceeds the threshold value for terminating theoperation of said despreader and thereafter storing the correspondingphase state; said control means being further responsive to a secondcondition wherein the difference value does not exceed the thresholdvalue, after operating said despreader over the range of phase states,for storing the phase state corresponding to a largest difference value;and where the stored phase state is used by said control means foradjusting the phase of the spreading code associated with the secondforward channel based on the first and second measures of receivedsignal energy.